This invention relates to apparatus for controlling a current type inverter adapted to drive an AC motor at variable speed, and more particularly to a controlling apparatus for controlling the output current vector of the current type inverter.
A well known construction of a current type inverter for controlling the operation of an AC motor is shown in FIG. 1. The inverter of the conventional construction converts DC current passing through a smoothing reactor 2 into three-phase AC current, which is supplied to the three phase windings R, S, and T of an AC motor 3. The inverter 1 comprises thyristors S.sub.1 through S.sub.6, diodes D.sub.1 through D.sub.6, and commutating capacitors C.sub.1 through C.sub.6, and is ordinarily controlled such that either one of the thyristors S.sub.1 to S.sub.3, and either one of the thyristors S.sub.4 to S.sub.6 are rendered conductive simultaneously. The vector of the current supplied from the inverter 1 to the AC motor 3 can occupy only six positions from I to VI as shown in FIG. 2.
In FIG. 2, the R phase winding of the AC motor 3 is taken as a reference axis. In a case where the thyristors S.sub.1 and S.sub.5 conduct, electric currents flow through the R and S phase windings of the electric motor 3. Since the R, S, and T phase windings of the same motor are spaced apart by an electric angle of 120.degree., the vector position of the synthesized current of the R phase and S phase currents is represented by I. Likewise, the vector positions of the synthesized currents flowing through other consecutive pairs of the phase windings are represented by II to VI, respectively.
In FIG. 3, there is illustrated a well known frequency controlling circuit of the inverter wherein each thyristor conducts for an electrical angle of 120.degree.. In this circuit, a reference numeral 4 designates a voltage-frequency converter which delivers a series of pulses P.sub.f, the frequency of which is varied in response to the magnitude of an input signal e.sub.f, and a reference numeral 5 designates a comparator which detects the polarity of the input signal e.sub.f. When the input signal e.sub.f is positive, the comparator 5 delivers an output e.sub.p in a "1" level, and when the input signal e.sub.f is negative, the comparator 5 delivers an output e.sub.p in a "0" level. The series of pulses P.sub.f generated from the voltage-frequency converter 4 are transferred by a transfer switch 6 controlled by the output e.sub.p of the comparator 5, so that the series of pulses P.sub.f are applied to up-input P.sub.u of a senary up-down counter 7 when the input signal e.sub.f is positive, and the same pulses P.sub.f are applied to down-input P.sub.D of the same counter 7 when the input signal e.sub.f is negative. The senary up-down counter 7 has six output positions e.sub.I through e.sub.VI, which are sequentially turned to "1" in an order of from e.sub.I to e.sub.VI when the output pulses P.sub.f are applied to the up-input P.sub.u of the counter 7. Conversely, when the pulses P.sub.f are applied to the down-input P.sub.D , one of the outputs e.sub.I through e.sub.VI becomes "1" sequentially in the order of from e.sub.VI to e.sub.I.
Reference numeral 8 designates a gate signal distributing circuit which delivers gate signals g.sub.1 through g.sub.6 to be applied to the thyristors S.sub.1 through S.sub.6. When two of the gate signals g.sub.1 through g.sub.6 selectively are turned to "1", two of the thyristors S.sub.1 through S.sub.6 (in FIG. 1) corresponding thereto are made conductive. The gate signal distributing circuit 8 may be preferably composed of six OR gates 81 through 86, and by such a construction, the output signals e.sub.I through e.sub.VI of the up-down counter 7 are caused to respectively correspond to the vector positions I through VI of the current to be supplied to the AC motor 3 as shown in FIG. 2.
FIG. 4 indicates a time chart for the operation of the device shown in FIG. 3 when a frequency control input signal e.sub.f of a positive constant value is applied thereto. A series of pulses P.sub.f delivered from the voltage-frequency converter 4 and having a frequency corresponding to the magnitude of the input signal e.sub.f, are controlled by the output e.sub.p of the comparator 5 delivered in response to the polarity of the input signal e.sub.f, such that the pulses P.sub.f are applied to the up-input P.sub.u or the down-input P.sub.D of the up-down counter 7 depending upon the polarity of the input signal e.sub.f. Each time when an up-pulse P.sub.u is applied, the up-down counter 7 shifts the position of the output "1" in the sequence of e.sub.I .fwdarw.e.sub.II .fwdarw. . . . e.sub.VI .fwdarw.e.sub.I.
According to the output condition of the up-down counter 7, the gate signal distributing circuit 8 selects the thyristors to be ignited, and the current vector supplied to the electric motor 3 is thereby stepwisely shifted in the sequence of I.fwdarw.II.fwdarw. . . . VI.fwdarw.I as shown in FIG. 2. This stepwise shift of the current vector is effectuated by the commutation of the inverter 1 in accordance with the timing by which the output signals of the up-down counter 7 and the gate signal distributing circuit 8 are varied by the output pulses P.sub.f of the voltage-frequency converter 4.
In a case where the input signal e.sub.f representing the frequency information is of a negative polarity, the output pulses P.sub.f of the voltage-frequency converter 4 are applied to the down-input P.sub.D of the up-down counter 7, and therefore the output signal "1" of the up-down counter 7 is shifted in the sequence of e.sub.VI .fwdarw.e.sub.V .fwdarw. . . . .fwdarw.e.sub.I .fwdarw.e.sub.VI. The current vector shown in FIG. 2 is thus rotated in the direction of VI.fwdarw.V.fwdarw. . . . .fwdarw.I.fwdarw.VI. Since the rotation fo current vector completes after six stepwise shifts, the one cycle period of the output pulses P.sub.f of the voltage-frequency converter 4 corresponds to 60.degree..
In the above described conventional arrangement where each of the thyristors conducts for 120.degree. continuously, the current vector supplied to the AC motor 3 is rotated stepwisely. Thus, a torque-ripple having a frequency proportional to the running speed of the AC motor is created in the torque of the same motor.
Although the torque-ripple causes no problem when the running speed is high, the one cycle period thereof becomes longer when the running speed of the AC motor is reduced, thus hampering smooth running of the AC motor.
In order to obviate the above described difficulty in an reduced speed operation of the AC motor, a device for modulating the pulse width (PWM) of the inverter 1 has been proposed to improve the waveform of the output current.
FIG. 5 illustrates an example of a previously proposed PWM controlling apparatus. In the apparatus, reference numerals 4 through 8 designate components similar to those described in FIG. 3. The apparatus further comprises a saw-tooth wave generator 9 for generating saw-tooth wave of a repetition frequency variable in accordance with the magnitude of the input signal e.sub.f, an oscillator 10 for generating a triangular waveform of a constant repetition frequency, a subtractor 11, and a comparator 12 for delivering an output signal e.sub.c which is "1" when the input signal is positive and is "0" when the input signal is negative. The apparatus further comprises transfer switches 13 through 18 which are interlinked with each other to be operated simultaneously. That is, when the control signal e.sub.c is "1", the transfer switches 13 through 18 are held in the indicated positions, and when the control signal e.sub.c is "0", the transfer switches are transferred to the opposite positions.
It is so arranged that when the frequency controlling input signal e.sub.f is applied to the saw-tooth wave generator 9, it delivers a saw-tooth wave e.sub.D having a repetition period corresponding to 60.degree. of the inverter operation and an amplitude equal to the amplitude of the triangular wave e.sub.T generated from the oscillator 10.
In FIG. 5, pulse signals a.sub.1 through a.sub.6 having a pulse width of 60.degree. as in the case of FIG. 3 are obtained at the outputs of the up-down counter 7. Thus, if the transfer switches 13 through 18 were held in the positions shown, it is apparent that the apparatus operates in the same manner at that of FIG. 3. However, when the transfer switches 13 to 18 are operated by the output e.sub.c, a PWM control can be obtained as described hereinafter with reference to the time chart shown in FIG. 6.
As is apparent from FIG. 6, the output pulse from the up-down counter 7 is shifted in the sequence of a.sub.1 .fwdarw.a.sub.2 .fwdarw.a.sub.3 .fwdarw. . . . each time an up-pulse P.sub.u is applied. On the other hand, a saw-tooth wave e.sub.D from the generator 9, which is in synchronism with the output pulses P.sub.f of the voltage-frequency converter 4, is subtracted in a subtractor 11 from the triangular wave e.sub.T delivered from the oscillator 10. A comparator 12 is supervising the variation of the polarity of the output signal of the subtractor 11, and transfers the movable contacts of the transfer switches from 13 to 18 each time when the polarity is changed. More specifically, when an instantaneous amplitude of the saw-tooth wave e.sub.D is smaller than that of the triangular wave e.sub.T (see FIG. 6), the transfer switches 13 to 18 are held at the upper contacting positions shown in FIG. 5, while when an instantaneous amplitude of the saw-tooth wave e.sub.D is greater than that of the triangular wave e.sub.T, the transfer switches are transferred to the lower contacting positions.
It is apparent that the transferring cycle period of the transfer switches is equal to the repetition period of the triangular wave e.sub.T, and the ratio between the actual periods where the transfer switches stay on the upper positions and the lower positions is determined by the magnitude of the saw-tooth wave e.sub.D. During a period corresponding to 60.degree. of the running frequency of the inverter wherein the pulse a.sub.1 is delivered from the up-down counter 7, the pulse a.sub.1 is caused to alternate between the output positions e.sub.I and e.sub.II by the operation of the transfer switch 13. In a range where the amplitude of the saw-tooth wave e.sub.D is smaller than that of the triangular wave e.sub.T, the pulse widths of the parts occupying the position e.sub.I are wide, and the pulse widths of the parts occupying the position e.sub.II are narrow. However, in accordance with the increase in amplitude of the saw-tooth wave e.sub.D, the widths of the first parts staying at the position e.sub.I are reduced, and the widths of the second parts staying at the position e.sub.II are increased as shown in FIG. 6.
In the next 60.degree. period where the output of the up-down counter 7 is changed from a.sub.1 to a.sub.2, the signal a.sub.2 is alternated between the output positions e.sub.II and e.sub.III by the operation of the transfer switch 14. The movement of the current vector for the AC motor 3 in the above described operation of the circuits will now be described with reference to FIG. 2.
During the time interval wherein the up-down counter 7 delivers an output a.sub.1, the corresponding current vector alternates between the positions I and II. In the case, when the amplitude of the saw-tooth wave e.sub.D is smaller than that of the triangular wave e.sub.T, the interval in which the current vector stays at the position I is longer than the period where the same vector stays at the position II. However, when the amplitude of the saw-tooth wave e.sub.D becomes large, the ratio of the time intervals of the vector staying at the positions I and II is gradually reversed. Moreover, during one cycle period of the triangular wave e.sub.T, the average position of the current vector, which is determined by the ratio of the intervals of the vector staying at the positions I and II, is rotated at a constant speed from the position I to the position II. The positions from II to VI of the current vector are also varied in the same manner as described above.
In the PWM control of the inverter, the current vector is rotated smoothly as described above, as the quick movements thereof determined by the frequency of the triangular wave e.sub.T are neglected. Thus, the torque ripple during the low-speed operation of the AC motor 3 can be eliminated, and a smooth running of the motor can be realized.
Although the PWM controlling apparatus has advantages as described above, it requires further circuit components such as the saw-tooth wave generator 9, triangular wave oscillator 10 and the like, thus increasing the manufacturing cost of the same apparatus.
Furthermore, as will be apparent from FIG. 6, the pulse widths of the output signals e.sub.I to e.sub.VI, which are applied to the gate signal distributing circuit 8, are made to be extremely narrow in the proximity of the minimum value and the maximum value of the saw-tooth wave e.sub.D. This pulse width corresponds to an interval between the two successive commutating instructions given to the inverter, and, in an ordinary inverter, the pulse width cannot be made smaller than a predetermined value. For instance, in the inverter 1 shown in FIG. 1, a certain interval of time is required from the instant of delivering a commutation instruction and to the instant of the completion of the commutation. If a following commutating instruction is given before the completion of the preceding commutation, there is a possibility of causing commutation failure. Thus, in the circuit shown in FIG. 5, the interval between the successive commutation instructions must be maintained at a longer value than the interval required for the completion of a commutation. Such a requirement further complicates the construction of the circuit.
In addition, if the above described restriction for the interval between successive commutation instructions is maintained, the ratio between the staying intervals of the current vector at two positions of the PWM control cannot be maintained at a desired value, and a low-frequency torque ripple would be created although the amount of the ripple itself is lower than that of the 120.degree. conduction operation of the inverter.
There has been proposed another PWM controlling apparatus for an inverter, wherein the waveform in one-cycle period of the output current is selected in accordance with a pattern as shown in FIG. 7, and specific higher harmonic components of the torque are thereby reduced to zero.
In FIG. 7, only the positive parts of the three phase output currents i.sub.R, i.sub.S, and i.sub.T are indicated. If an interval .alpha..sub.1 in FIG. 7(a) is assumed to be 12.degree., the fifth higher harmonic component becomes zero. Likewise, in FIG. 7(b), if an interval .alpha..sub.1 is assumed to be 5.82.degree. and .alpha..sub.2 is assumed to be 16.25.degree., the fifth and seventh harmonic components become zero. Furthermore, when the number of pulses in one-half period of the pattern is increased, higher harmonic components of higher orders can be reduced to zero.
In the above described example, however, from the requirement that the minimum commutation period must be selected in consideration of the commutation capability of the inverter 1, and that a minimum amount of higher harmonic components must be maintained, various patterns must be prepared for the operations of variable running frequencies of the inverter, so that a suitable pattern can be selected for each running frequency. This means that the procedure for preparing waveform patterns as shown in FIG. 7 is not suitable for the application to an inverter of variable frequency.
Thus, it is apparent that neither of the conventional PWM controlling apparatus is suitable because of their complicated construction, necessity of adjustments, and the high manufacturing costs.